The forward converter is a common circuit topology used to transform electric energy from a source at a given potential to a destination load at a different potential. A typical forward converter includes a transformer having a primary winding and at least one secondary winding. The primary winding of the transformer is coupled to a source of power, usually DC power, via a primary switch or transistor. The secondary winding is coupled to a load via an output rectifier circuit comprising two commutating diodes and an output filter. The primary switch generally comprises a semiconductor switching device such as a FET or bipolar-junction transistor (BJT). When the primary winding is energized by the closing of the primary switch, i.e., the ON-period of the switch, energy is immediately transferred to the secondary winding, hence the name forward converter. In a typical forward converter, energy is stored in the transformer magnetizing and leakage inductances during the ON-period of the switch. During the OFF-period, the voltage across the transformer primary winding reverses, stored energy is dissipated, and magnetic flux in the core is reset. It is necessary to limit the voltage generated during the OFF-period to avoid damage to the switch or transistor.
More particularly, during the conduction or ON period of the switching transistor, current is transferred from the primary DC power source through the transformer to the output circuit. During the OFF period of the switching transistor, the magnetizing current in the transformer is returned to the primary DC source, resetting the flux in the transformer core, prior to the next cycle of operation.
FIGS. 1 and 2 illustrate two versions of a conventional forward converter wherein a third winding of the converter's transformer is used to limit the maximum reset voltage and to reset the core of the transformer when the primary switch is opened. Forward converter 1000 is shown in the circuit of FIG. 1 and comprises a transformer 1010 having a primary winding 1020 and a secondary winding 1022, a resistor 1026 in parallel with a clamp capacitor 1025 connected in series with a diode 1030, all of which is connected in parallel across the primary winding 1020, an auxiliary circuit 1040 connected to a third winding 1045, and a primary FET switch 1050. On the secondary side of transformer 1010 are the two commutating diodes 1055, 1060 which are coupled to an inductor 1070 to provide output power (Vout) to a load 1072 and output capacitor 1074.
Resistor 1026 acts to dissipate the energy in clamp capacitor 1025. Clamp capacitor 1025 along with diode 1030 limit the maximum reset voltage across switch 1050. Auxiliary circuit 1040 provides a source of power Vcc to the converter's control circuit via a diode 1075 and a capacitor 1080. Power is coupled to the converter from an input power source (Vin) which is connected across the series combination of the primary winding 1020 and the primary switch 1050. The power dissipated by the clamp in forward converter 1000 is often 5% to 10% of the output power (Vout), which is too high to be wasted as heat in the clamp resistor 1026.
Another simpler method for limiting the voltage generated during the OFF-period of a power supply system is shown in the forward converter 2000 shown in FIG. 2. The forward converter 2000 includes a third winding 2022 but it is connected across the input supply Vin just by a diode 2030. In this example, the third winding usually has the same number of turns as the primary, which means that the peak voltage developed across the switching transistor 2050 during the OFF period is twice the primary DC supply voltage. For a nominal rectified line input of 300V DC, the peak switch voltage would therefore be 600V, requiring a switching transistor voltage rating of at least 700V in such an example. Note that the maximum permissible conduction period or duty cycle of switching transistor 2050 is usually 50% of the total cycle time, to allow time for the transformer flux to be reset during the OFF period and avoid core saturation.
The above mentioned clamp winding methods improve the efficiency of a power converter, but they have disadvantages such as: a) the clamp voltage is proportional to, and increases with, the input voltage; b) the clamp winding can not be perfectly coupled to the primary winding, and so is unable to clamp all of the energy; and c) the high frequency oscillations between the coupling inductance and stray capacitances is a source of EMI.
It is also known in the art to employ other clamping methods to improve the efficiency of a power converter, such as the active clamp, but these methods do not completely resolve the foregoing disadvantages. See, for example, U.S. Pat. No. 4,441,146 wherein the third winding is eliminated and replaced by a series combination of a storage capacitor and an auxiliary switch coupled across either the primary or secondary winding. The auxiliary switch is operated counter to the primary switch, i.e., it is open when the primary switch is closed and closed when the primary switch is open.
Utilizing auxiliary converters to power a control circuit is also known. For instance, power for a control circuit is normally taken from a winding on the power transformer or from the input supply via an auxiliary converter. This auxiliary converter may be a linear regulator, which has been proven to be very inefficient, or a free running switching converter such as a flyback or blocking oscillator. A typical auxiliary converter, such as a free running converter, can generate a wide range of low frequency oscillations, resulting in EMI, due to the varying beat frequency between the main and auxiliary converters. Such a free-running converter is often unacceptable for certain telecommunication applications, such as telephone exchanges. The disadvantage of using these types of auxiliary converters is that it necessitates the costly and time-consuming task of synchronizing the auxiliary converter to the main converter.
Representative devices that utilize various clamping arrangements combined with converters to protect switching devices from high voltage transients and to recover the energy stored in the clamp during the OFF-period by using a switching means are described as follows: U.S. Pat. No. 4,607,322 to Henderson discloses an energy recovery snubber that includes a push-pull converter and clamp capacitor that returns energy to a supply by a second switch and by the windings on a transformer; U.S. Pat. No. 4,286,314 to Molyneux-Berry relates to an inverter circuit for minimizing switching power losses comprising a switch, clamp capacitor, and a second switch connected to an inductor coupled to a clamp capacitor, to resonantly discharge energy for each cycle, while returning that energy to the supply input; U.S. Pat. No. 4,438,486 to Ferraro for a low loss snubber for power converters involves a clamp capacitor, energy retrieved via separate (synchronized) flyback converter, where the FET in the flyback detects a higher voltage than the clamp, and experiences switching losses; U.S. Pat. No. 4,912,620 to O'Dell relates to a lossless clipper with peak regulation feature which comprises a forward converter and clamp capacitor, and a buck converter for returning clamped energy to a supply source, with optional regulation of the clamped voltage. The circuits described in the above-mentioned patents are illustrated in FIGS. 3-6, whereby transient energy is utilized, rather than wasted, in the OFF period of a power supply system.
In FIG. 3, a snubber circuit 3000 is shown comprising a switch transistor 3010 connected to primary windings 3050, 3060 of the main transformer 3090. The transistor 3010 is operated between ON and OFF states by a switch control (not shown) so that alternating current power is developed in the secondary winding (not shown) of the transformer 3090. The snubber circuits 3020, 3030 are connected in series to the auxiliary primary winding 3040, which are inductively linked to the primary windings 3050,3060.
The snubber circuit 3020 in FIG. 3 includes a capacitor 3080, which is coupled to a switching transistor 3010 of the power converter. The switching transistor 3010 modulates the flow of current through the primary windings 3040, 3050, 3060 of the transformer 3090. The auxiliary primary winding 3040 is coupled by means of a second snubber switch 3030 to the junction between the first snubber switch 3020 and the capacitor 3080. Shortly before turning off the switching transistor 3010, the first snubber switch 3020 is closed and the second snubber circuit 3030 is opened so that the snubber capacitor 3080 is connected in parallel with the switching transistor 3010. When the power transistor is turned off, the transient energy of the primary winding 3060 of the transformer 3090 is stored by the snubber capacitor 3080. Then, the first snubber circuit 3020 is opened and the second snubber switch 3030 is closed to couple the transient energy to the auxiliary primary winding 3040 to possibly drive a load coupled to a secondary winding of the transformer 3090.
In FIG. 4, a known inverter circuit 4000 for minimizing switching power losses is shown. The circuit 4000 comprises a switch 4040 connected in series to a load 4035 across a transformer. A first diode 4050 and a clamp capacitor 4045 are serially connected so as to be in shunt with the switch 4040. A second diode 4055, inductor 4030, a second switch 4060 and the capacitor 4045 are connected in series. A power converter circuit 4070 consists of a transformer 4010 having a primary winding 4020 and secondary winding 4025 across the load 4035. Hence, the inverter circuit 4000 has a switch 4040 which is conductive periodically to provide pulse power from a power source 4065 to a load 4035. Diode 4050 and capacitor 4045 clamp the voltage of switch 4040. Second switch 4060, diode 4055 and inductor 4030 return the energy in the clamp capacitor 4045 back to the power source 4065.
In FIG. 5, a low loss snubber for power converter 5000 using a flyback converter is shown. The snubber 5000 comprises a capacitor 5045 in series with a diode 5050 and a energy retrieval circuit 5065. The capacitor 5045 and diode 5050 are connected in series across the first switch 5035. The diode 5050 is poled to conduct current to charge the capacitor 5045 when the first switch 5035 is turned off. The energy retrieval converter 5065 comprises a transformer 5010 having a primary winding 5020 and a secondary winding 5025, and a second switch means 5030 that is repeatedly turned off and when the first switch 5035 is in a closed position. An inductor means or the primary winding 5020 of the transformer 5010 is in series with the second switch 5030, and the capacitor 5045 transfers the energy stored in the inductor means 5020 to an external load when the second switch 5030 is opened.
In FIG. 6, a snubber lossless clipper circuit 6000 having a peak regulation feature is shown. The clipper circuit 6000 limits or clamps the flyback voltage pulses generated by a transformer 6070 having a primary winding 6020 and a secondary winding 6010 arranged across a switching transistor 6030. The snubber circuit 6000 dissipates no energy. When the transistor is turned off, the snubber circuit 6000 modulates the magnitude of the flyback pulses to thereby modulate the voltage across the transformer 6070. In the snubber circuit 6000, when the transistor 6030 is turned on, the current flows through the primary winding 6020 to switch the transistor 6030 to ground, which causes the current to be delivered to the secondary winding 6010 of the transformer 6070 to a choke inductor 6080 and to a load 6085. When the transistor 6030 is turned off, the voltage at the lead 6090 begins to raise. The snubber or clipper circuit 6000 coupled to the transformer 6070 prevents the voltage at the lead 6090 from increasing beyond a value which could damage the transistor 6030.
Although the above-described known approaches for clamping a converter's reset voltage and recovering the energy stored in the clamp during the transistor OFF period have created some efficiencies in returning energy back to the input power supply, other problems remain. One problem with switches 3030, 4060, 5030 and 6065 in FIGS. 3-6 is that they all turn on with voltage across them, i.e., they are all hard switching devices, which generates switching noise and EMI, and creates switching power losses. The other problem with prior art converters is that if a third winding on the main power transformer were used to provide auxiliary power, it wouldn't function when the main converter is shut off. The transformer in FIG. 5, or another winding added to the inductors 4030 or 6060 would only give auxiliary power when the main converter is running. The prior art solution is therefore to include an additional auxiliary power converter that runs continuously to ensure the availability of auxiliary power when needed.
There is therefore a need in the power conversion art for a means of efficiently clamping a forward converter reset voltage with minimum losses in energy and minimum EMI, and that can also provide auxiliary power even when the main converter is shut down.